Power supply device and semiconductor device

ABSTRACT

Switching loss is reduced by decreasing the switching frequency of a PFC power supply in light load condition, whereas the switching frequency is maintained high in heavy load operation. Efficiency in light load operation is thus improved without enlarging a boosting inductor and an output smoothing capacitor. A capacitor is provided in a triangular wave generating circuit and the triangular wave generating circuit outputs a triangular wave by charging and discharging this capacitor. Charging and discharging of the capacitor are controlled by an oscillation frequency control circuit output current which is input to a comparator.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.13/052,409, filed Mar. 21, 2011, which claims priority to JapanesePatent Application No. 2010-95915 filed, Apr. 19, 2010, the disclosureof which, including the specification, drawings and abstract, isincorporated herein by reference in its entirety.

BACKGROUND

The present invention relates to a power supply device and asemiconductor device for use therein and, particularly, to a powersupply device having a power factor correction circuit.

Power supply devices for use in servers, home appliances, etc. have aproblem in which reactive power is generated due to a difference betweeninput voltage and input current waveforms, when converting alternatecurrent (AC) to direct current (DC). Another problem concerned is thatsuch a difference between the waveforms of an input voltage and an inputcurrent gives rise to a harmonic current that becomes noise havingeffect on other electrical equipment and the like. As a countermeasureagainst this problem, power factor correction (PFC) is performed thatshapes the waveform of an input current into a phase and waveformequivalent to those of an input voltage.

Examples of power factor correction circuits are described in JapaneseUnexamined Patent Publication No. Hei 10(1998)-174428 (Patent Document1), Japanese Unexamined Patent Publication No. 2009-261042 (PatentDocument 2), and Japanese Unexamined Patent Publication No. 2007-195282(Patent Document 3).

RELATED ART DOCUMENTS Patent Documents

-   [Patent Document 1] Japanese Unexamined Patent Publication No. Hei    10(1998)-174428-   [Patent Document 2] Japanese Unexamined Patent Publication No.    2009-261042-   [Patent Document 3] Japanese Unexamined Patent Publication No.    2007-195282

SUMMARY

In a power supply device, when a MOSFET or the like used as a switch isturned ON/OFF, changes in a gate drive current, a drain voltage, and adrain current change result in a loss (switching loss). This gives riseto a problem of a decrease in power conversion efficiency. The lower theswitching frequency, that is, the smaller the number of times ofswitching per unit time, the switching loss becomes smaller. However,simply reducing the number of times of switching per unit time is notpractical, because it is required for this purpose to enlarge aninductor and an output smoothing capacitor of the power supply device,which leads to an increase in a die area or a packaging area.

One object of the present invention is to provide means for improvingthe characteristics of a power supply device having a power factorcorrection circuit and a semiconductor device for use therein.

The above-noted and other objects and novel features of the presentinvention will become apparent from the following description in thepresent specification and the accompanying drawings.

Typical aspects of the invention disclosed in this application aresummarized as follows.

A power supply device pertaining to a typical embodiment of the presentinvention is characterized by having a PFC controller that includes anerror amplifier amplifying a difference between an output voltage of avoltage diving circuit which divides a voltage on an output terminal anda first reference voltage, an oscillation frequency control circuitconverting an output of the error amplifier to a current, a triangularwave generating circuit outputting a triangular wave from an outputcurrent of the oscillation frequency control circuit, and a PWM controlcircuit opening and closing a switch, based on the triangular waveoutput by the triangular wave generating circuit.

The power supply device may be characterized in that the triangular wavegenerating circuit includes a capacitor and outputs the triangular waveby charging and discharging the capacitor.

The power supply device may be characterized in that the triangular wavegenerating circuit further includes a first current mirror circuithaving two outputs and a second current mirror circuit having oneoutput, a current output of the second current mirror circuit is areplica of one current output of the first current mirror circuit, theother current output of the first current mirror circuit and the currentoutput of the second current mirror circuit are coupled via a MOSFET,and the MOSFET conducts charging and discharging the capacitor coupledto a node coupling the MOSFET and the current output of the secondcurrent mirror circuit.

The power supply device may be characterized in that the triangular wavegenerating circuit further includes a comparator to which a constantvoltage is input, the comparator compares a voltage on the node couplingthe capacitor, the current output of the second current mirror circuit,and the MOSFET to this constant voltage, and the capacitor is chargedand discharged by opening and closing the MOSFET.

The power supply device may be characterized in that a voltage on thenode coupling the current output of the second current mirror circuit,the MOSFET, and the comparator is output as the triangular wave.

The power supply device may be characterized in that the oscillationfrequency control circuit includes a transistor and an output of theerror amplifier is input to a gate terminal of the transistor.

The power supply device may be characterized in that a third currentmirror circuit is coupled to a collector terminal of the transistor anda current output of the third current mirror circuit is an output of theoscillation frequency control circuit.

The power supply device may be characterized in that the oscillationfrequency control circuit further includes a constant current source, afourth current mirror circuit is coupled to the collector terminal ofthe transistor, and a current resulting from subtracting an output ofthe fourth current mirror circuit from an output current of the constantcurrent source is an output current of the oscillation frequency controlcircuit.

The power supply device may be characterized in that the oscillationfrequency control circuit further includes a clamp circuit coupled to anemitter terminal of the transistor.

The power supply device may be characterized in that the power supplydevice is a continuous mode type in which currents are interleaved andthe PWM control circuit controls a switch in each of subsystemsrespectively.

A power supply device pertaining to a typical embodiment of the presentinvention is characterized by having a resistor for output currentmeasurement and a PFC controller that includes a potential differenceamplifier amplifying a voltage across the resistor for output currentmeasurement, a triangular wave generating circuit outputting atriangular wave from an output current of the potential differenceamplifier, and a PWM control circuit opening and closing a switch, basedon the triangular wave output by the triangular wave generating circuit.

It is possible to improve the characteristics of a power semiconductordevice for use therein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram representing a configuration of a powersupply device pertaining to a first embodiment of the present invention.

FIG. 2 is a circuit diagram representing a configuration of a PWMcontrol circuit.

FIG. 3 is a circuit diagram representing a concrete configurationcomprehending an oscillation frequency control circuit and a triangularwave generating circuit pertaining to the first embodiment of thepresent invention.

FIG. 4 is a chart to explain operation of the triangular wave generatingcircuit pertaining to the first embodiment of the present invention.

FIG. 5 is a graph to explain how the frequency of the oscillationfrequency control circuit changes.

FIG. 6 is a graph showing a relationship among load factor, outputvoltage of an error amplifier, and switching frequency.

FIG. 7 is a circuit diagram representing a concrete configurationcomprehending an oscillation frequency control circuit and a triangularwave generating circuit pertaining to a second embodiment of the presentinvention.

FIG. 8 is a graph to explain how the frequency of the oscillationfrequency control circuit changes, pertaining to the second embodimentof the present invention.

FIG. 9 is a graph showing the waveforms of outputs of a PFC controllerin which the oscillation frequency control circuit pertaining to thesecond embodiment of the present invention is applied.

FIG. 10 is a circuit diagram representing a configuration of a powersupply device pertaining to a third embodiment of the present invention.

FIG. 11 is a circuit diagram representing a configuration of a powersupply device pertaining to a fourth embodiment of the presentinvention.

DETAILED DESCRIPTION

As for embodiments, as will be described hereinafter, a plurality ofsections or embodiments is separately described, as necessary for thesake of expedience. Unless otherwise specified, however, these sectionsor embodiments are not independent of each other; rather, they correlatein such a manner that one of them provides an example of modification toor further detailed or supplementary explanation of another one of themin part or in whole. In embodiments, as will be discussed hereinafter,where the number of elements or the like (including the number ofpieces, a value, a quantity, a range, etc.) is mentioned, unlessotherwise specified and unless it is obviously limited to a given numberin principle, it is not limited to the given number and may be eithermore than or less than the given number.

Moreover, it goes without saying that components of an embodiment, aswill be discussed hereinafter, are not always indispensable ones, unlessotherwise specified and unless a component is obviously considered asindispensable in principle. Although not restrictive, circuit elementsmaking up each function block of an embodiment are formed over asemiconductor substrate like monocrystalline silicon by integratedcircuit technology such as CMOS (Complementary Metal Oxide SemiconductorField Effect Transistor). MOSFET (Metal Oxide Semiconductor Field EffectTransistor), if mentioned in an embodiment, does not exclude a non-oxidefilm as a gate insulation film. A gate length and a gate width of aMOSFET are expressed with symbols L and W, respectively. W/L denotes aratio of a gate width to a gate length.

In the following, embodiments of the present invention will be describedusing the drawings.

First Embodiment

<<Overview of Power Supply Device>> FIG. 1 is a circuit diagramrepresenting a configuration of a power supply device pertaining to afirst embodiment with a load Ld that is coupled to this power supplydevice.

This power supply device includes a PFC controller 10 (a PFC controllerIC, a power factor correction controller, a power factor correctioncontroller IC, a semiconductor device) and a switch Q100, besides adiode bridge B1, an inductor L1 (a boosting inductor, a coil, a boostingcoil), a diode D1, a voltage dividing circuit 1, an output smoothingcapacitor Cout, and a current detecting resistor Rs, and a resistor RAC.The power supply device that takes a power factor correction measureusing such PFC controller is also called an active filter type. Althoughthe diode bridge B1 is a full-wave rectifying circuit in which fourdiodes are coupled in a bridge form, its simplified form is shown as inFIG. 1.

The inductor L1 is used to boost an input voltage (voltage output fromthe diode bridge B1). To the inductor L1, the switch Q100 is coupled. Asthe switch Q100, an element such as a MOSFET or IGBT (Insulated GateBipolar Transistor) is used. The diode D1 is a passive element forrectification to control current to follow in one direction.

The voltage dividing circuit 1 is comprised of resistors Rf1 and Rf2.This circuit divides a potential difference between an voltage Vout onan output terminal and a ground level and outputs a result as voltageinformation VFB (=Vout Rf2/(Rf1+Rf2)) to the PFC controller 10.

The current detecting resistor Rs is a grounded resistor to detect avoltage (current information output VCS) between the diode bridge B1 andthe ground potential.

The current detecting resistor RAC is a resistor to detect currentoutput information IAC. One end of the current detecting resistor RAC iscoupled to a node coupling the diode bridge B1 and the inductor L1 andthe other end thereof is input to a multiplier #1 in a PWM controlcircuit 10-4.

Current detection information IAC is a voltage that is determined by aresistance value of the current detecting resistor RAC and a potentialdifference between both ends of the current detecting resistor RAC.

The PFC controller 10 is a control circuit that takes inputs of thevoltage information VFB and the current information output VCS andperforms switching of the switch Q100.

The PFC controller 10 in the present first embodiment includes an erroramplifier 10-1, an oscillation frequency control circuit 10-2, atriangular wave generating circuit 10-3, and a PWM control circuit 10-4.Although not restrictive, these circuits making up the PFC controller 10is formed within a single semiconductor chip (semiconductor substrate).

The error amplifier 10-1 is an operational amplifier that amplifies adifference between an output voltage (voltage information VFB) of thevoltage dividing circuit 1 and a reference voltage VS1. If we let Adenote the gain of the operational amplifier, then the output voltage V1of the error amplifier 10-1 is expressed as follows: V1=A×(VS1−VFB). Ina case where the load Ld coupled to the power supply device is heavy (inother words, power consumed by the load Ld is large), the output voltageVout decreases and VFB becomes low. Hence, V1 becomes high. Conversely,in a case where the load coupled to the power supply device is light (inother words, power consumed by the load Ld is small), V1 becomes low.That is, the output voltage V1 of the error amplifier 10-1 is changeddepending on the load Ld. Here, one feature is that the output voltageV1 of the error amplifier 10-1 continuously changes depending on theload Ld.

The oscillation frequency control circuit 10-2 is a control circuit thatadjusts the output of a constant current source incorporated therein,based on the output voltage V1 of the error amplifier 10-1, and controlsthe triangular wave generating circuit 10-3.

The triangular wave generating circuit 10-3 is a circuit that outputs atriangular wave (ramp wave) voltage VT to the PWM control circuit 10-4.

The PWM control circuit 10-4 is a control circuit that changes the pulsewidth of a signal to be input to a gate terminal of the switch Q100 inorder to make the switch Q100 On (conducting)/Off (non-conducting).

When the switch Q100 is On, energy is stored in the inductor L1; when itis Off, the input voltage and the energy in the inductor L1 are added.Then, this energy is supplied as a current to the output smoothingcapacitor Cout via the diode D1. Thereby, it is possible to boost theinput voltage. In addition, by adjusting the On/Off ratio (duty ratio)of the switch Q100, it is possible to adjust a value of the outputvoltage Vout.

<<PWM Control Circuit>> FIG. 2 is a circuit diagram representing aconfiguration of this PWM control circuit 10-4.

The PWM control circuit 10-4 includes a multiplier #1, an erroramplifier #2, a comparator #3, a flip-flop #4, a resistor #5, and abuffer #6.

The multiplier #1 is an analog multiplier that multiplies the outputvoltage V1 of the error amplifier 10-1 by the output voltage of thecurrent detecting resistor RAC and outputs an voltage proportional tothe product.

The error amplifier #2 amplifies an error between the output of themultiplier and the ground voltage. At this time, via the resistor #5,the current information output VCS is coupled to the output of themultiplier which is input to the error amplifier #2. Thereby, a feedbackdepending on a voltage that is applied to the resistor #5 via theresistor Rs is input to the error amplifier #2.

The comparator #3 compares an output voltage of the error amplifier #2and a triangular wave voltage VT which is output by the triangular wavegenerating circuit 10-3 and outputs an error between them.

The flip-flop #4 is an RS flip-flop circuit for driving an On/Offcontrol terminal (gate terminal) of the switch Q100. When an outputvalue of “1” of the above comparator #3 is input to a set terminal ofthe flip-flop #4, the flip-flop #4 outputs “1”. When a reset pulse isinput to a reset terminal of the flip-flop #4 (i.e., the reset terminalturns to “1”), the flip-flop #4 outputs “0”.

The buffer #6 is an analog buffer circuit that outputs a linear outputvoltage Vgd in response to a predefined range of input voltages, so thatits output voltage can drive the On/Off control terminal of the switchQ100. This output voltage Vgd continuously changes depending on theoutput voltage V1 of the error amplifier 10-1.

<<Oscillation Frequency Control Circuit and Triangular Wave GeneratingCircuit>> FIG. 3 is a circuit diagram representing a concreteconfiguration comprehending the oscillation frequency control circuit10-2 and the triangular wave generating circuit 10-3 pertaining to thefirst embodiment.

The oscillation frequency control circuit 10-2 is comprised of aresistor R4, a transistor Qr, and P-channel type MOSFETs Qp1 and Qp2making up a current mirror circuit CM1.

The transistor Qr provides an emitter-follower output. The transistor Qradjusts a current flowing across the resistor R4 and a reference currentIref according to a voltage (i.e., the output voltage of the erroramplifier 10-1) that is applied to its gate terminal. As the transistorQr, a bipolar transistor or MOSFET is used.

The current mirror circuit CM1 reflects the reference current Irefflowing in a collector terminal of the transistor Qp1 to a drainterminal of the transistor MOSFET Qp2, thus making an output current I1.At this time, a ratio between the currents Iref and I1 is determined bythe W/L ratio of each of the MOSFETs Qp1 and Qp2.

The triangular wave generating circuit 10-3 includes a current mirrorcircuit CM2 and a current mirror circuit CM3 for drawing an inputcurrent (I1 in FIG. 3) into a main operating part of the triangular wavegenerating circuit 10-3.

The current mirror circuit CM2 is comprised of N-channel type MOSFETsQn1, Qn2, Qn3. The current mirror circuit CM3 is comprised of P-channeltype MOSFETs Qp3, Qp4.

To a drain terminal of a MOSFET Qn1 in the current mirror circuit CM2,the output current I1 of the oscillation frequency control circuit 10-2is input. Depending on this current I1, a current I2 and a current I3flow through a MOSFET Qn2 and a MOSFET Qn3, respectively. A ratiobetween the currents I1 and I2 is determined by the W/L ratio of each ofthe MOSFETs Qn1 and Qn2. Similarly, a ratio between the currents I1 andI3 is determined by the W/L ratio of each of the MOSFETs Qn1 and Qn3.Further, depending on the above current I2, a current I4 flows through aMOSFET Qp4. A ratio between the currents I2 and I4 is determined by theW/L ratio of each of the MOSFETs Qp3 and Qp4.

A drain terminal of the MOSFET Qp4 and a drain terminal of the MOSFETQp3 are electrically coupled via a MOSFET Qsw1. A voltage on a node #Acoupling the MOSFET Qp4 and the MOSFET Qsw1 is input to a comparatorCmp. A capacitor C1 is also coupled to the coupling node #A. As will bedetailed further on, by repeating storing charge on the capacitor C1 anddischarging, the voltage on the coupling node #A is changed and thetriangular wave voltage VT is generated.

The MOSFET Qsw1 is a switch to turn on/off the electrical couplingbetween the drain terminal of the MOSFET Qp4 and the MOSFET Qn3. Anoutput voltage Vc of the comparator Cmp, which will be described later,is input to a gate terminal of the MOSFET Qsw1.

When the MOSFET Qsw1 is Off (in the non-conducting state), most of thecurrent I4 flowing through the MOSFET Qp4 is directed toward thecapacitor C1. This is because the comparator Cmp and other circuitscoupled to the coupling node #A are regarded as having a high impedance.Therefore, if we let Ic denote a current flowing into the capacitor C1,a relation that I4 is approximately equal to Ic becomes true. Due tothis current Ic, charge is stored in the capacitor C1 and the voltage(triangular wave voltage VT) on the coupling node #A rises.

On the other hand, when the MOSFET Qsw1 is On (in the conducting state),the charge stored in the capacitor C1 flows toward ground GND via theMOSFETs Qsw1 and Qn3. If we let I4 denote a current flowing from theMOSFET Qp4 to the coupling node #A, Id denote a current flowing from thecapacitor C1 to the coupling node #A, and I3 denote a current flowingthrough the MOSFETs Qsw1 and Qn3, a relation: Id=I3−I4 holds true. Here,because the value of the current I4 does not become 0 in the currentmirror circuit CM3, the value of the current I3 needs to be larger thanthe value of the current I4 to discharge the charge stored in thecapacitor C1. By way of example, let us consider a case in which the W/Lratios of the MOSFETs Qn1, Qn2, Qp3, and Qp4 are set so that the valuesof the currents I1, I2, and I4 are substantially equal. In this case, bymaking the W/L ratio of the MOSFET Qn3 larger than the W/L ratio of theMOSFET Qn1 and making the value of the current I3 larger than thecurrent I1, the value of the current I3 can be made larger than thevalue of the current I4.

That is, by making the current I3>the current I4, the charge stored inthe capacitor C1 is discharged; accordingly, the potential (triangularwave voltage VT) on the coupling node #A falls.

A voltage dividing circuit Div which is comprised of resistors R1, R2,R3 generates a reference voltage Vct for the comparator Cmp.

An N-channel type MOSFET Qsw2 is a switch to determine whether or notthe resistors R2 and R3 are coupled in parallel. When the MOSFET Qsw2 isOff (in the non-conducting state), the reference voltage Vct becomes avalue of VctH yielded by diving a power supply voltage VDD by theresistors R1 and R2: i.e., VctH=VDD×R2/(R1+R2). On the other hand, theMOSFET Qsw2 becomes On (transits into the conducting state), theresistors R2 and R3 are coupled in parallel and the value of thereference voltage Vct falls to VctL=VDD×R2 R3/(R1+R2) (R2+R3). Here, asan example of the power supply voltage VDD, approximately 5 V is used.

The comparator Cmp is for comparing the reference voltage Vct and thevoltage on the coupling node #A. The comparator Cmp outputs a Low levelsignal (a voltage of 0) when the voltage on the coupling node #A islower than the reference voltage Vct and outputs a High level signal (avoltage of VcH) when the voltage on the coupling node #A is higher thanthe reference voltage Vct. Depending on the output voltage Vc of thecomparator Cmp, turning the MOSFET Qsw1 and the MOSFET Qsw2 On/Off iscontrolled.

<<Operation of Triangular Wave Generating Circuit>> Using FIG. 3 andFIG. 4, how the voltage (triangular wave voltage VT) on the couplingnode #A changes is explained (let us consider a case when, as an initialstate, the MOSFETs Qsw1 and Qsw2 are Off and the reference voltage Vctis VctH as mentioned above).

(1) When the MOSFET Qsw1 is Off, the current Ic flows in the capacitorC1 and charge is stored therein; due to this, the voltage on thecoupling node #A rises (for a period of Tr).

(2) When the voltage on the coupling node #A has reached the abovereference voltage VctH, the output voltage Vc of the comparator Cmpturns to High (voltage VcH).

(3) When the output voltage Vc of the comparator Cmp turns to High, theMOSFET Qsw1 turns On, charge is discharged from the capacitor C1, andthe voltage on the coupling node #A falls. Further, the MOSFET Qsw2turns On and the reference voltage Vct falls to VctL (for a period ofTf).

(4) When the voltage on the coupling node #A has fallen down to thereference voltage VctL, the output voltage Vc of the comparator Cmpturns to Low (voltage of 0).

(5) When the output voltage Vc of the comparator Cmp turns to Low, theMOSFETs Qsw1 and Qsw2 turn Off and the reference voltage Vct starts torise to VctH again.

By repeating the foregoing sequence from (1) to (5), the voltage on thecoupling node #A has a triangular waveform, as shown in FIG. 4( a).

<<Dependency of Switching Frequency on Load Factor>> FIG. 5 is a chartshowing the waveforms of (a) output voltage Vct of the error amplifier10-1, (b) reference voltage Vct of the comparator Cmp, (c) outputvoltage Vc of the comparator Cmp, (d) triangular wave voltage VT(voltage on the coupling node #A in FIG. 3), and (e) output voltage Vgdof the PWM control circuit 10-4, when the load Ld in FIG. 1 is (1)light, (2) middle, and (3) heavy.

Three patterns are explained separately, when the power supply device isin (1) light load operation, (2) middle load operation, and (3) heavyload operation.

(1) As described above, when in light load operation, the output voltageV1 of the error amplifier 10-1 is a low value. Therefore, the value ofthe current Iref in FIG. 3 is small; accordingly, the values of thecurrents I1 to I4 are small. Due to this, time taken to store charge inthe capacitor C1 and discharging time become longer and, thus, theperiods Tr and Tf in FIG. 4 become longer. Hence, the frequency of theoutput voltage VT of the triangular wave generating circuit 10-3 is low.Accordingly, the frequency of the output voltage Vgd of the PWM controlcircuit 10-4 as a signal that controls turning the switch Q100 On/Off islow. That is, the switching frequency of the switch Q100 is low. Here,light load refers to a load factor on the order of 20%, by way ofexample.

In contrast, when in (3) heavy load operation, the output voltage V1 ofthe error amplifier 10-1 is a high value and, therefore, the switchingfrequency of the switch Q100 is high, contrary to (1) light loadoperation. When in (2) middle load operation, the switching frequency ofthe switch Q100 is middle between that in (1) light load operation andthat in (3) heavy load operation (3). Here, by way of example, theswitching frequency may be on the order of 30 KHz in (1) light loadoperation, 50 KHz in middle load operation (2), and 100 kHz in heavyload operation (3).

As previously noted, the output voltage V1 of the error amplifier 10-1continuously changes depending on the load Ld. Moreover, since theoutput voltage Vgd of the PWM control circuit 10-4 continuously changesdepending on the output voltage V1 of the error amplifier 10-1, theoutput voltage Vgd of the PWM control circuit 10-4 continuously changesdepending on the load Ld. Hence, it is possible to continuously changethe frequency of the output voltage Vgd of the PWM control circuit 10-4as a signal that controls turning the switch Q100 On/Off (switchingfrequency) depending on the load Ld.

By configuring the power supply device in the way described above,circuit operation as will be described further below is feasible.

FIG. 6 is a graph showing a relationship among load factor of the powersupply device, output voltage V1 of the error amplifier 10-1, andswitching frequency. Here, the load factor is a ratio of a momentaryload to a maximum rated load of the power supply device; in other words,a ratio of a momentary power to a maximum rated power of the powersupply device. It may also refer to a ratio of the load Ld in FIG. 1 tothe maximum rated power of the power supply device. Exemplary values ofthe maximum rated power may be on the order of 500 W for the powersupply device for a personal computer and on the order of 2000 W for thepower supply device for an air-conditioner. Exemplary values of loads in(1) light load operation, (2) middle load operation, and (3) heavy loadoperation are, for example, given approximately as follows. For thepower supply device for a personal computer; in (1), load ranging from 0to 150 W and load factor ranging from 0 to 30%; in (2), load rangingfrom 150 W to 350 W and load factor ranging from 30% to 70%; and in (3),load ranging from 350 W to 500 W and load factor ranging from 70% to100%. For the power supply device for an air-conditioner; in (1), loadranging from 0 to 500 W and load factor ranging from 0 to 25%; in (2),load ranging from 500 W to 1000 W and load factor ranging from 25% to50%; in (3), load ranging from 1000 W to 2000 W and load factor rangingfrom 50% to 100%.

In the graph, a solid line (a) represents a relationship between loadfactor and switching frequency with regard to the present firstembodiment. For the sake of reference, a chain line (b) represents arelationship between load factor and switching frequency in the case ofa power supply device described in Patent Document 1.

In the present first embodiment, the output voltage V1 from the erroramplifier 10-1 linearly changes depending on the load factor. That is,it can be regarded that what amount of the load coupled to the powersupply device is determined by the error amplifier 10-1 and,correspondingly, the error amplifier outputs a suitable value of theoutput voltage V1. Thereby, it is possible to linearly change theswitching frequency, so that the switching frequency will be low whenthe power supply device is in light load operation and will be high whenin heavy load operation.

By contrast, in the power supply device described in Patent Document 1,the switching frequency can assume only two values and the switchingfrequency does not linearly change depending on the load factor.

If the power supply device having the PFC controller of the presentfirst embodiment is used, it is possible to maintain the switchingfrequency high during heavy load operation. For this reason, theinductor L1 and the output smoothing capacitor Cout are not required tohave large values of inductance and capacitance. Hence, it is avoidableto increase the die area or packaging area of the power supply device.It is also possible to decrease the switching frequency of the switchQ100 in light load condition. For this reason, it is possible to improvethe power conversion efficiency in light load operation. Therefore, itis possible to achieve a high power conversion efficiency over a widerange of loads by improving the power conversion efficiency in lightload operation, which has been a problem in the past.

If the power supply device having the PFC controller of the presentfirst embodiment is used for a home appliance product such as, e.g., anair-conditioner, it is possible to improve the power conversionefficiency during steady load operation in addition to improving thepower conversion efficiency at maximum power required when the homeappliance product is powered on.

This power supply device also satisfies standards, inter alia, ENERGYSTAR (a power saving program for electrical equipment promoted by the USEnvironmental Protection Agency) and 80plus (a power saving program forelectrical equipment promoted by the 80plus program (www.80plus.org)).

Second Embodiment

Then, a second embodiment is described using the drawings.

FIG. 7 is a circuit diagram representing a concrete configurationcomprehending an oscillation frequency control circuit and a triangularwave generating circuit pertaining to the second embodiment.

A basic configuration of the present second embodiment is the same asfor the first embodiment, as shown in FIG. 1. Among the components, theerror amplifier 10-1, the triangular wave generating circuit 10-3, andthe PWM control circuit 10-4 are the same as for the first embodiment.Therefore, further descriptions are provided hereinafter only for anoscillation frequency control circuit 10-2 b.

The oscillation frequency control circuit 10-2 b includes, besides atransistor Q11, a current mirror circuit CM4 including MOSFETS Q12 ANDQ13, a constant current source IS1, resistors R5, R6, a transistor Q14,and a reference voltage VS2.

Output of the error amplifier 10-1 is input to a gate terminal of thetransistor Q11, as is the case for the first embodiment.

To an emitter terminal of the transistor Q11, the resistors R5, R6 whichact as a load resistance are coupled. The transistor Q11 provides anemitter-follower output.

The current mirror circuit CM4 replicates a current flowing in a drainterminal of a MOSFET Q13 to a drain terminal of a MOSFET Q12.

The constant current source IS1 is a source from which a constantcurrent I11 flows. This current I11, a current (designated by I12herein) flowing in the current mirror circuit CM4, and an output currentI13 of the oscillation frequency control circuit 10-2 b have thefollowing relation.

I11=I12+I13

This equation is transformed into:

I13=I11−I12

From this equation, it can be understood that the smaller the current tobe input to the current mirror circuit CM4, the larger will be thecurrent to be input to the triangular wave generating circuit 10-3.

Then, a clamp circuit comprised of the reference voltage VS2 and thetransistor Q14 is described.

In this clamp circuit, an emitter terminal of the transistor Q14 iscoupled to a node coupling the resistors R5 and R6. Consequently, if avoltage on the emitter terminal of the transistor Q11 is higher than avoltage on the emitter terminal of the transistor Q14, no current flowsin the current mirror circuit CM4. That is, when the output of the erroramplifier 10-1 is equal to or more than a voltage prescribed by theclamp circuit, the triangular wave output of the triangular wavegenerating circuit 10-3 and the oscillation frequency of the PWM controlcircuit 10-4 remain constant. Conversely, after the output of the erroramplifier 10-1 becomes lower than the voltage prescribed by the clampcircuit, the oscillation frequency of the PWM control circuit 10-4starts to fall.

Then, operation is explained using a graph.

FIG. 8 is a graph to explain how the frequency of the oscillationfrequency control circuit 10-2 b changes, pertaining to the secondembodiment. FIG. 9 is a graph showing the waveforms of outputs of thePFC controller 10 in which the oscillation frequency control circuit10-2 b pertaining to the second embodiment is applied.

As can be seen in FIG. 8, in the oscillation frequency control circuit10-2 b of the preset second embodiment, the output voltage of the erroramplifier 10-1 increases linearly, until the load reaches a certainvalue. The triangular wave output of the triangular wave generatingcircuit 10-3 and the oscillation frequency of the PWM control circuit10-4 also increase proportionally.

However, when the output of the error amplifier 10-1 reaches and exceedsthe voltage predefined by the clamp circuit in FIG. 7 (corresponding toload (2) in FIG. 8), the triangular wave output of the triangular wavegenerating circuit 10-3 and the oscillation frequency of the PWM controlcircuit 10-4 no longer increase (see FIG. 8 and FIG. 9).

This is clearly shown with regard to the triangular wave output of thetriangular wave generating circuit 10-3, when you see a load range (2)and (3) in FIG. 8. That is, it is evident that the triangular wavevoltage VT of the triangular wave generating circuit 10-3 and theoscillation frequency of the PWM control circuit 10-4 do not change,when the output of the error amplifier 10-1 becomes equal to or morethan a certain level.

Thereby, it is possible to improve switching loss without increasing thefrequency of the switch Q100 above the certain level.

Third Embodiment

Then, a third embodiment is described.

In the present third embodiment, the invention is envisioned to beapplied to a continuous mode, interleaving type PFC controller. Here,the continuous mode is a mode in which switch operation is performed ina state that duplicate inductor currents are allowed to flow. Theinterleaving type is a type in which a pair of switches is operatedalternately.

FIG. 10 is a circuit diagram representing a configuration of a powersupply device pertaining to the third embodiment.

The power supply device in the present third embodiment has twosubsystems, in each of which charging and discharging an inductor areconducted by a switch, because of the interleaving type. One subsystemis comprised of an inductor L1, a diode D1 and a switch Q100 and theother subsystem is comprised of an inductor L2, a diode D2, and a switchQ200.

The PWM control circuit 10-4 is replaced by a PWM control circuit 10-4 cto fulfill its role to control the two switches, i.e., one more switchadded.

The PWM control circuit 10-4 c takes input of voltage information aboutthe switch Q100 and the switch Q200 in addition to input signals to thePWM control circuit 10-4. In order to detect voltage information foreach switch, a resistor is inserted between each switch and ground. Asshown in FIG. 10, a resistor Rs1 is inserted in one subsystem includingthe switch Q100 and a resistor Rs2 is inserted in the other subsystemincluding the switch Q200, respectively. Resistance values of theseresistors are assumed to be Rs1=Rs2; in this case, there is no need tobias the control within the PWM control circuit 10-4 c. However, theresistance values of the resistors Rs1 and Rs2 may be biased for anypurpose.

Based on the voltage information for each switch, the PWM controlcircuit 10-4 c is capable of implementing control in response to thestatus of each switch.

The PWM control circuit 10-4 c has dual outputs in order to control theswitches Q100 and Q200.

By configuring the power supply device in the way described above, it ispossible to decrease the switching frequency and improve switching losseven in the power supply device using the continuous mode, interleavingtype PFC controller.

Fourth Embodiment

A forth embodiment is described. The present fourth embodiment ischaracterized in that a signal serving as a reference for control is acurrent that is supplied to an output terminal, instead of a voltageprovided by the voltage dividing circuit 1.

FIG. 11 is a circuit diagram representing a configuration of a powersupply device pertaining to the fourth embodiment.

In the fourth embodiment, a resistor for output current measurement R1is inserted just before the output terminal. In the previousembodiments, the output of the error amplifier 10-1 is input to theoscillation frequency control circuit 10-2 (10-2 b is also applicable).In contrast, in the present fourth embodiment, a potential differenceamplifier 10-5 d is added. A feature of the fourth embodiment is asfollows. A voltage across the resistor for output current measurement R1is amplified by the potential difference amplifier 10-5 d and input tothe oscillation frequency control circuit 10-2. Thereby, control of thePFC controller 10 is implemented, based on the current flowing acrossthe resistor for output current measurement R1.

By configuring the power supply device in this way, it is also possibleto decrease the switching frequency.

Although the technical concept of the present fourth embodiment isapplied to the first embodiment, it can also be applied to the secondembodiment.

While the invention made by the present inventors has been describedspecifically based on its embodiments hereinbefore, it will be obviousthat the present invention is not limited to the described embodimentsand various modifications may be made without departing from the scopeof the invention.

What is claimed is:
 1. A method of controlling a power supply deviceconfigured to supply DC power to a load circuit (Ld), the power supplydevice having an inductor (L1) configured to receive AC rectified power;and a switch transistor (Q100) coupled with the inductor and switchingcurrent through the inductor on and off, the method comprising the stepsof: (a) generating a voltage information signal (VFB) based on an outputvoltage (Vout) applied to the load circuit; (b) generating a firstvoltage signal (V1) based on the voltage information signal (VFB); and(c) generating a PWM signal (Vgd) based on the first signal (V1), thePWM signal (Vgd) controlling the switch transistor, wherein a frequencyof the PWM signal (Vgd) continuously increases as the output voltage(Vout) increases.
 2. The method according to claim 1, wherein thefrequency of the PWM signal (Vgd) linearly increases as the outputvoltage (Vout) increases.
 3. The method according to claim 1, wherein,in the step (a), the voltage information signal (VFB) is generated bydividing the output voltage (Vout) using two resistors (Rf1/Rf2).
 4. Themethod according to claim 1, wherein, in the step (b), the voltageinformation signal (VFB) and a reference voltage (VS1) are inputted toan error amplifier (10-1) and the first signal (V1) is generated by theerror amplifier (10-1).
 5. The method according to claim 1, wherein, inthe step (c), the PWM signal (Vgd) is generated based on a triangularwave signal, wherein the triangular wave signal changes depending on thefirst signal (V1).
 6. The method according to claim 4, wherein thetriangular wave signal is generated by charging and discharging acapacitor.
 7. A method of controlling a power supply device configuredto supply DC power to a load circuit (Ld), the power supply deviceincluding: an output terminal and a ground terminal; a diode bridge (B1)to which the alternate current voltage is inputted; an inductor (L1)coupled to an output of the diode bridge; a diode (D1) whose anode iselectrically coupled to the inductor and whose cathode is electricallycoupled to the output terminal; a capacitor (Cout) electrically coupledbetween the output terminal and the ground terminal; a switch transistor(Q100) electrically coupled between the anode of the diode and theground terminal, a voltage dividing circuit (1) coupled with the outputterminal and the ground terminal; and an error amplifier (10-1) coupledwith the voltage dividing circuit and a reference voltage (VS1); anoscillation frequency control circuit (10-2) coupled with the erroramplifier; a triangular wave generating circuit (10-3) coupled with theoscillation frequency control circuit; and a PWM control circuit (10-4)coupled with the triangular wave generating circuit and the, the methodcomprising the steps of: (a) generating a voltage information signal(VFB) in the voltage dividing circuit by dividing a voltage between theoutput terminal and the ground terminal; (b) generating a first voltagesignal (V1) in the error amplifier (10-1) by amplifying a differencebetween the voltage information signal (VFB) and the reference voltage(VS1); (c) converting the first voltage signal (V1) to a first currentsignal in the oscillation frequency control circuit (10-2); (d)generating a triangular wave signal (VT) based on the first currentsignal; (e) generating a PWM signal (Vgd) based on the triangular wavesignal (VT) in the triangular wave generating circuit (10-3); and (f)controlling the switch transistor (Q100) with the PWM signal (Vgd). 8.The method according to claim 7, wherein a frequency of the PWM signal(Vgd) continuously increases as the output voltage (Vout) increases. 9.The method according to claim 8, wherein the frequency of the PWM signal(Vgd) linearly increases as the output voltage (Vout) increases.
 10. Themethod according to claim 7, wherein, in the step (e), the triangularwave signal is generated by charging and discharging a capacitor. 11.The method according to claim 7, wherein the triangular wave generatingcircuit further comprises a first current mirror circuit having twooutputs, and a second current mirror circuit having one output, whereina current output of the second current mirror circuit is a replica ofone current output of the first current mirror circuit, wherein theother current output of the first current mirror circuit and the currentoutput of the second current mirror circuit are coupled via a MOSFET,and wherein the MOSFET conducts charging and discharging the capacitorcoupled to a node coupling the MOSFET and the current output of thesecond current mirror circuit.
 12. The method according to claim 11,wherein the triangular wave generating circuit further comprises anoperational amplifier to which a constant voltage is inputted, andwherein the operational amplifier compares a voltage on the nodecoupling the capacitor, the current output of the second current mirrorcircuit, and the MOSFET to the constant voltage, and the capacitor ischarged and discharged by opening and closing the MOSFET.